Method and System for Transmitting Signals with Reduced Spurious Emissions

ABSTRACT

An RF power amplifier architecture minimizes spurious emissions by breaking the transmitted signal into narrow spectrum sub-bands, amplifying each separately, and then combining the signals for transmission purposes.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.10/861,476 filed Jun. 7, 2004, which claims priority under 35 USC 119(e)to provisional patent application No. 60/477,155 filed on Jun. 10, 2003,the disclosure of each of which is incorporated by reference in itsentirety.

STATEMENT AS TO RIGHTS TO INVENTIONS MADE UNDER FEDERALLY SPONSOREDRESEARCH OR DEVELOPMENT

This invention was made with Government support under ContractF3062-03-C-0141 awarded by the Air Force. The Government has certainrights in this invention.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention is directed to an RF power amplifier architecturethat significantly reduces an amplifier's out-of-band emissions.

2. Background Art

A principle goal of communication systems is to maximize spectrumefficiency via the use of broadband waveforms transmitting overnon-contiguous spectrum and to minimize the waveform's adjacent powerlevel. A goal of the Defense Advanced Research Projects Agency (DARPA)in Future Combat Systems (FCS) and other programs is the development ofthe orthogonal frequency-division modulation (OFDM) for use withtactical systems. The OFDM waveform has perhaps the best combination ofmultipath and Eb/No properties of any waveform. Unfortunately, the OFDMwaveform has a high peak to average voltage ratio, which requires veryhigh amplifier linearity to suppress out-of-band emissions. This is avery significant problem that prevents the OFDM waveform in somescenarios because high power, linear amplifiers are not cost effectivein many designs. Some designers believe that the OFDM out-of-bandemission problem is so severe that single carrier waveforms with specialequalization are the preferred solution (see Falconer, David et al.,Frequency Domain Equalization for Single-Carrier Broadband WirelessSystems, IEEE Communications Magazine, April 2002). The proposedtechnology directly mitigates the OFDM out-of-band emission problem,thus enabling OFDM to be widely applicable in tactical situations.

Minimizing out-of-band and spurious emissions is a very challengingaspect of multi-band radio design. FCC regulations and interferencepredictions (detailed below) present typical maximum emission values. Toachieve low emissions, a combination of high third order interceptamplifiers and tracking filters are required. The filters must be high-Qbandpass filters with the passband set close to the desired outputbandwidth. The filters must have high third order intercept points toavoid contributing to the problem they are fixing. These amplifiers andfilters are expensive, are large, require high prime power, and areheavy. State-of-the-art broadband, high performance tunable filters aremanufactured by PoleZero Corporation. The high IP₃ performance (>+50dBm) PoleZero product is the “Power-Pole” Filter. This device requiresapproximately 7.5 W of prime power, is several inches in size, and isexpensive (˜$2 k each). The frequency coverage is only 10 MHz to 700 MHzand each device has a 3:1 tuning ratio.

The required amplifier performance level to obtain low spuriousemissions is difficult if not impossible to easily attain in prior artradios. Many radios economize on this part of the design and sufferserious operational limitations or have great difficulties in gettingspectrum authorization.

These problems are compounded in the next generation of multi-bandradios because the large frequency range and the use of variabletransmit bandwidths increases the number of required filters. The goalof transmitting a non-contiguous spectrum requires even greater filterflexibility to accommodate “tailoring” of the transmitted spectrum. Itis believed that a “brute-force” amplifier/filtering approach to achieveacceptably low spurious emissions with a multi-band radio, and toachieve reasonable cost, size, weight, and power goals is not possiblewith current technology.

The maximum permitted spurious transmitted power levels are quite smalland require high RF performance to achieve. The maximum power can bedetermined via two methods. Both methods are used in the current debateon authorizing Ultra-Wide Band (UWB) devices (see filed comments onDocket 98-153 at www.fcc.gov). This debate is relevant because broadbandwaveforms create wide bandwidth noise over the approximately the samefrequency band that is widely used for terrestrial, tacticalcommunications (20 MHz to 3,000 MHz). Many users within this band havefiled comments regarding wideband noise interference applicable to theirspecific systems, making available detailed, applicable interferenceanalysis.

The first method (“Noise Floor Method”) to estimate the maximum spurioustransmitter power level determines the maximum broadband noise levelthat can be transmitted that would cause a small (3 dB) rise in thevictim receiver's thermal noise floor at a certain distance. Using thefree-space range equation, omniantennas and a 6 dB victim noise figure,FIG. 13 shows the maximum transmitted noise level in dBm/MHz versusfrequency for ranges of 10 meters, 100 meters, and 1000 meters. In mostcases, frequency sharing using broadband waveforms will occur withdistant primary users at 1000 m range or greater. An exception is theGPS band, which will be heavily used in close proximity (a few meters)to tactical receivers.

The second method (“Part 15 Method”) to estimate the maximum spurioustransmitted power level is to adopt FCC Part 15 (CFR Part 15.209),standards for the amount of unintended power radiated. This standard isa field strength level (100 uV/meter from 30-88 MHz, 150 uV/meter from88 216 MHz, 200 uV/meter from 216-960 MHz, and 500 uV/meter at greaterthan 960 MHz) at 3 meters range from the device. FIG. 13 shows thislevel as “Part 15” transformed back to transmitted power using thefree-space range equation and omni-antenna gains. Above 960 MHz, theallowed transmission power is −41.2 dBm/MHz.

These two methods provide different power levels, which is partially thekey issue in the UWB debate and also creates uncertainty on what levelsshould be adopted. Several groups have argued that the Part 15 levelsare too high for UWB devices and need to be reduced, especially in theGPS bands (1559 MHz-1610 MHz). The FCC has proposed a 12 dB reduction tothe Part 15 limit to a value of −53.2 dBm/MHz, FCC Notice of ProposedRulemaking FCC 00-163. These arguments are related to the high peak toaverage power ratio of the UWB signal and the large number of expectedUWB devices, and may not be applicable to an advanced broadbandwaveform. The UWB arguments are also focused on interference with nearby(several meters) devices and not with distant (100s of meters) victimreceivers. It is believed that the Part 15 requirements are more thanadequate for tactical purposes and that the Noise Floor Method is themost applicable. The exception is within the GPS bands wherein the FCC'srecommendation of the maximum emission level of −53.2 dBm/MHz appears tobe reasonable. Table 1 summarizes the recommended emission levelsdetailed above.

TABLE 1 Allowable Band Emissions Allowable Emission Allowable EmissionLevel in a 1 MHz Level in a 61 kHz Band Bandwidth Bandwidth Comment 30MHz to 100 MHz −45 dBm/MHz −57.1 dBm “Noise Floor Method” 100 MHz to −35dBm/MHz −47.1 dBm “Noise Floor Method” 1,000 MHz 1,000 MHz to −35dBm/MHz −47.1 dBm “Noise Floor Method” with 3,000 MHz¹ 20 dB margin toaccount for directional antennas 1559 MHz- −53.2 dBm/MHz   −65.3 dBm GPSband - Follow FCC 1610 MHz recommendation ¹Except for GPS band.

What these maximum permitted spurious power levels means is that anon-contiguous waveform must have very low out-of-band emissions toavoid causing interference, and a need exists to have systems that arecapable of achieving these low out-of-band emissions.

Present day amplifiers are incapable of achieving these low emissions.In this regard, estimates of the out-band emission levels of RFamplifiers with different third order intercept points (IP₃) are shownin FIGS. 14A-C. Shown are typical output spectrums with a 20 MHztransmitted waveform with a 5 MHz spectrum gap, 1 W output, and anamplifier IP₃ value of +40 dBm (FIG. 14A), +50 dBm (FIG. 14B), and +60dBm (FIG. 14C). The “noise” level within the spectrum gap (at 100 MHz)is −20 dBm (IP₃=+40 dBm) (FIG. 14A), −40 dBm (IP₃=+50 dBm) (FIG. 14B),and −60 dBm (IP₃=+60 dBm) (FIG. 14C).

Comparing with the allowable emission levels as shown in Table 1indicates that with this waveform and power level, an output amplifierwith IP₃˜+55 dBm is required. An amplifier with IP₃>+60 dBm is requiredto protect the GPS band, which is not feasible within the SWPconstraints of tactical radios. Current multi-band amplifiers have IP₃values of about +42 dBm, Stanford Microdevices SGA-9289 (IP₃=42.5 dBm).Thus, burdensome post amplifier bandpass filtering will be required toachieve low spurious output levels with this example waveform, which webelieve is similar to what is envisioned for future tactical radios.

Commercially available multi-band RF amplifiers have limited dynamicrange performance. It is likely that commercial amplifiers small enoughfor tactical operations will not have high enough IP₃ performance toproduce a useful spectrum notch.

Table 2 shows the performance of several types of commercially availableRF amplifiers. The performance of the recently introduced WJCommunications AH1 amplifier is shown in the first row. The WJ AH1 issmall enough to be useful for small, handheld devices. It's IP₃ level of+41 dBm would produce power out-of-band emissions similar to what isshown in FIG. 14A. The Mini-Circuits ZHL-42W is a connectorizedinstrumentation that covers about the frequency range of interest (30MHz to 3,000 MHz), but also has a low IP₃ value. The Mini-CircuitsZHL-5W-1 has higher output power and IP₃ value, but doesn't cover theupper part of the desired frequency range. This is probably due to straycapacitance effects in the baluns and transformers used in the design.

An extreme example is the Spectran MCPA 4080, which is a narrowband PCSbase station amplifier. It has excellent RF performance over a narrow(1930 MHz to 1990 MHz) frequency range and would support a lowout-of-band emission, non-contiguous waveform. But the required powerlevel of 1130 W is prohibitive, even for vehicle applications. Thus,current amplifiers are still lacking in supporting a low out-of-bandemission, non-contiguous waveform.

TABLE 2 Performance comparison of several amplifier types. FrequencyOutput Bias Amplifier Type Range Power Power IP₃ Efficiency WJCommunications MMIC 250 MHz to +21 dBm 2 W 41 dBm 6% AH1 component 3000MHz Mini-Circuits - Connectorized 10 MHz- +28 dBm 12.5 W 38 dBm 5%ZHL-42W instrumentation 4200 MHz amplifier Mini-Circuits - Connectorized1 MHz- +37 dBm 79 W 45 dBm 6% ZHL-5W-1 instrumentation 500 MHz amplifierAmplifier Research Rack mounted 800 MHz to +39 dBm 150 W 43 dBm 5% 5S1G4instrumentation 4,200 MHz amplifier Spectran MCPA 4080 Rack mounted,1930 MHz to +49 dBm 1130 W 81 dBm 7% single band 1990 MHz PCS basestation amplifier

As such, there is a need for improved amplifiers that reduce anamplifier's spurious emission to maximize spectrum efficiency. Thepresent invention responds to this need by the development of animproved amplifier system that reduces out-of-band emissions, thiseliminating significant interference to existing narrow bandwidth users.

SUMMARY OF THE INVENTION

It is a first object of the present invention to provide an improvedamplifier system to reduce an amplifier's out-of-band emissions.

The invention entails a new and novel RF power amplifier architecturethat significantly reduces an amplifier's out-of-band emissions. Thiswill enable a principle communications goal, to maximize spectrumefficiency via the use of broadband waveforms transmitting overnon-contiguous spectrum. The proposed amplifier design reducesout-of-band emissions in the waveform's spectrum holes so they will notcause significant interference to existing narrow bandwidth usersoperating within the waveform holes. Existing amplifiers will createsignificant out-band emissions and will severely limit the use ofnon-contiguous waveforms. The invention also reduces the normal adjacentchannel out-of-band emissions.

The proposed sub-band transmitter (SBT) concept minimizes spuriousemissions by breaking the transmitted signal into narrow spectrumsub-bands, amplifying each separately, and then combining the signals.This reduces distortion because the spurious spectrum width is dependenton the input spectrum width to each amplifier. By amplifying thespectrum in narrow sub-bands, the spurious energy is not allowed tospread to frequencies away from the desired signal.

The SBT concept's benefit is that it reduces the required amplifier IP₃value (saving prime power) and mitigates the need for highly flexibletracking filters. The only transmit filtering required will be for theremoval of harmonic distortion, which can be accomplished with a simplefilter design. One embodiment uses four sub-bands to obtain >40 of dB ofspurious signal reduction. Of course, other embodiments could use morethan four sub-bands, e.g., up to ten sub-bands, which will be practicaland will provide exceptional RF performance requiring the use of minimalguard bands.

The inventive sub-band transmitter can easily be implemented. Futuretransceivers use digital methods and D/A converters at the modem, andcan easily divide the signal into spectral sub-bands. Having parallelup-converter chains is easily accommodated because the local oscillatorcan be shared between the channels and the remaining “parallelized”parts are low cost and low power.

Other objects and advantages of the present invention will becomeapparent as a description thereof proceeds.

In satisfaction of the foregoing objects and advantages, the presentinvention provides improvements in methods and systems involvingtransmission of signal by vastly reducing spurious emissions. In onemethod aspect, the invention is an improvement in the method oftransmitting a signal based on a received analog input signal whereinthe input signal is amplified for transmission via an antenna. Accordingto the invention, the input signal is subdivided into a plurality ofsub-bands, with each subdivided signal up-converted, amplified, and thencombined for transmission via the antenna.

The invention also entails calibrating each sub-band's relativeamplitude and phase values so that the combined signals approximate theinput signal to be transmitted. One calibration method involves makingan initial relative amplitude and phase measurement, and then makingsubsequent amplitude only measurements using a swept CW signal for eachsub-band to determine amplitude and phase corrections for each sub-band.

An alternate calibration method involves generating a pair of CW signalsand iteratively varying the relative amplitude and phases of the pair ofCW signals until a summed output is minimized to determine the relativeamplitude and phase values for the subdividing step.

A third alternative involves generating a test signal, amplifying thetest signal and looping back the amplified signal to a receiver andmodulator for measuring of the amplitude and phase of the looped backsignal relative to the amplitude and phase of a signal corresponding tothe analog input. A variation of the third alternative involvesproviding a test signal, amplifying the test signal and transmitting theamplified test signal using a first transceiver to a second transceiver.Then, the relative amplitude and phase values of the transmitted testsignal is determined by digitizing the test signal, transforming thedigitized signal into sub-bands, and amplifying the sub-bands. Thesecond transceiver transmits the determined relative amplitude and phasevalues back to the first transceiver to set the amplitude and phase ofthe sub-bands for the subdividing step. The first receiver can include aloop back switch to allow transmission of the test signal to the secondtransceiver and receipt of the determined relative amplitude and phase.The second transceiver can include a loop back switch to receive thetransmitted test signal, and to permit transmission of the determinedrelative amplitude and phase of the sub-bands to the first transceiver.The test signal can comprise a number of test signals, each test signaltransmitted over the same frequency range using different spreadingcodes.

The invention also provides a system architecture that is improved oversystems wherein an analog input signal is transmitted via an antennausing an up-converter and an amplifier. According to the invention, aprocessing module is provided that converts the received analog inputsignal into a digital signal, parses the digitized signal into aplurality of sub-bands, calibrates the sub-bands, and converts eachsub-band into an analog signal. An up-converter and amplifier isprovided for each sub-band. A combiner combines each of the amplifiedsub bands for transmission. Means can be provided for calibrating outamplitude and phase differences in each digitized sub-band. Thecalibrating means can include means for generating a test signalcorresponding to each sub-band and measuring the output signal for eachsub-band to determine amplitude only or amplitude and phase corrections.

In an alternate mode, the system calibrates using means for generating apair of test signals and varying the amplitude and phase of the testsignals, and means for detecting the zero output signal of the combinerto determine the amplitude and phase values for the digitized sub-bands.

In yet another alternate mode, the calibrating means further comprisesmeans for generating a test signal for output by one of the amplifiers,and a loop back switch downstream of the one amplifier, a transceiverreceiving output from the loop back switch; output of the transceiverbeing sent to the processing module, the processing module measuring theamplitude and phase of the output from the transceiver relative to thetest signal input to determine amplitude and phase corrections. Afurther variation of this mode, first and second transmitter systems areprovided. The output of the loop back switch of the first transmittersystem is transmitted to the transceiver of the second transmittersystem, the second transmitter system transforming the test signal intosub-bands and numerically determining the relative amplitude and phasesof the sub-bands, and then transmitting the numerically determinedrelative amplitude and phases to the first transmitter system forsetting of the amplitude and phase values.

The invention also entails a universal processing module fortransmitting an analog input signal that comprises an analog to digitalconverter for digitizing the analog input signal and means forsubdividing the digitized signal into a plurality of sub-bands andcalibrating the sub-bands to account for amplitude and phasedifferences. A digital to analog converter is provided for eachsub-band, the output of each digital to analog converter adapted forlater up-converting, amplification, and combining into one signal fortransmission purposes. A detector is also provided that is adapted toreceive a sample of the combined output signal of the amplifiedsub-bands, and to send the sample to the subdividing means as part ofcalibrating the sub-bands. The subdividing means further comprises amicroprocessor adapted to receive the output from the detector forcalibration purposes.

BRIEF DESCRIPTION OF THE DRAWINGS

Reference is now made to the drawings of the invention wherein:

FIG. 1 is a schematic of one embodiment of a sub-band transmitter (SBT)system that reduces the out-of-band emissions by using multipleamplifier chains to transmit spectrum sub-bands;

FIG. 2 is a schematic of a prior art transmit amplifier system with asingle signal path that amplifies the entire signal spectrum;

FIG. 3 is a schematic of an alternate transmitter system that usesmultiple amplifiers;

FIGS. 4A and 4B are schematics showing calibration methods usingamplification only amplifier only and amplification/phase control,respectively;

FIG. 5 is a schematic showing how the relative amplitude and phasebetween any two channels can be found by canceling the output signal;

FIG. 6 is a schematic showing a loop back calibration method;

FIG. 7 is a schematic showing a communications link loop backcalibration method;

FIGS. 8A and 8B show a waveform in normal operation with each channeltransmitting in different sub-bands, and a calibration waveform witheach channel transmitting on the same sub-band simultaneously,respectively;

FIG. 9 is a schematic showing a universal sub-band transmitterprocessing mode architecture;

FIG. 10 is a schematic showing current amplifier technology using afeed-forward method to subtract non-linear signal components;

FIGS. 11A-11C shows a first graph of the sub-band transmitter outputspectrum and four amplifiers, a second graph of the output spectrum witha single amplifier transmitting the entire spectrum (baseline), and athird graph showing the output spectrum with four amplifiers in paralleleach transmitting the entire spectrum (baseline with equivalentamplifiers), respectively;

FIGS. 12A-12C deal with a non-contiguous spectrum and show a first graphof the sub-band transmitter output spectrum and four amplifiers, asecond graph of the output spectrum with a single amplifier transmittingthe entire spectrum (baseline), and a third graph showing the outputspectrum with four amplifiers in parallel each transmitting the entirespectrum (baseline with equivalent amplifiers), respectively;

FIG. 13 is a prior art graph showing transmitted power density (dBm/MHz)versus frequency per FCC Part 15 and to equal the noise floor of a 6 dBNF receiver with omni-antennas at different ranges (10 m, 100 m, and1000 m); and

FIGS. 14A-14C are graphs showing output spectrum of a 20 MHz BWbroadband waveform (with a 5 MHz gap) at 1 W output with three differentamplifiers (IP₃=40 dBm top, IP₃=50 dBm middle, and IP₃=60 dBm bottom),respectively.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The inventive sub-band transmitter system greatly reduces spuriousemissions by amplifying narrow spectrum sub-band “slices” and thencombining them to obtain the full spectrum. FIG. 1 shows an exemplarysystem architecture 10 that uses multiple digital to analog (D/A)converters 1 that receive different waveforms or ‘sub-bands’ 3A-3D thatare produced by a modem 5. In this embodiment, the sub-band waveform'sbandwidth is approximately one quarter the final signal bandwidth, butthe bandwidth may vary depending on the number of sub-bands. Thesub-band signals 3A-3D are converted to analog using the D/A converters1, up-converted at 7, amplified at 9, using four amplifiers, and thencombined using combiner 11 before being sent to the antenna 13.

For comparison, FIG. 2 shows a conventional architecture with a singletransmitter chain 20, using a single path that converts the signal 21from the modem 22 to analog at 23, up-converts at 25, amplifies at 27,and then sends the signal to the antenna 29.

In the sub band method, amplitude and phase differences between thetransmitter channels are minimized during the system's manufacture, anddrifts with time or temperature are calibrated out. The calibrationtechnique (described later) utilizes a built in test system with a loopback circuit to the associated receiver, or via feedback from otherreceivers. The calibration is further simplified if the modem isflexible enough to put the entire signal through one channel or sendcontinuous wave (CW) calibration signals on each channel for diagnosticor calibration purposes (also described later).

An alternate configuration 30 shown in FIG. 3 (that will be analyzed inthe next section) has parallel amplifiers 31 that operate on the entiresignal spectrum 33, with the amplifier outputs 35 combined at 37 anddirected to the antenna 39. The parallel amplifiers provide a higherdegree of linearity or net higher third order intercept point, but aswill be shown in comparison to the invention below, provide marginalreduction in spurious emissions.

A key sub-band transmitter issue is calibration of each sub-band signalpath's relative amplitude and phase so that the combined signalapproximates the desired signal. The specific relative amplitude andphase requirements will vary for different waveform types, but arelikely to be a several tenths of a dB in amplitude and a few degrees inphase. These will probably not be met with RF hardware manufacturingtolerances and will probably require periodic recalibration to accountfor temperature effects, different output frequencies, or output powerlevels. The calibration may also need to be updated if there are changesin the output impedance due to changes in the antenna or the feed line.

The present invention has three alternate schemes of calibrations. Thefirst is to directly measure the transfer characteristics of eachchannel using a swept CW signal. In this mode, the signal does notrequire demodulation. The second mode is a cancellation technique wherethe relative amplitude and phase of pairs of signals are iterativelyvaried until the summed output is minimized, thus determining therelative amplitude and phase values. The third mode involves usingwaveforms with different spreading codes on each channel, demodulatingthe signals, and using signal processing to determine the relativeamplitude and phase.

The calibration method selection depends on the characteristics of thehardware's amplitude and phase differences between the sub-bands and thelevel of integration between the SBT amplifier and the receiver.

The first and simplest method involves initially measuring the relativeamplitude and phase of each sub-band using external equipment duringmanufacture and then an amplitude only calibration for in-fieldadjustments. The amplitude only system is shown in FIG. 4A as referencenumeral 40. The system 40 would use swept CW signals 41 generated by themodem 5 on each sub-band one at a time and then measure the output powerlevel 45 from amplifier 9 using a coupler 47 and power detector 49. Thedetector would be very simple and would not use any filtering or RFfrequency conversion. This method presumes that an amplitude onlymeasurement would be sufficient to determine both amplitude and phasecorrections. If the amplitude only measurement is insufficient todetermine both amplitude and phase correction, then an I/Q demodulator51 with a dual power detector 53 could be used to measure amplitude andphase as shown in FIG. 4B.

The principle advantage of this technique is that it is independent ofthe receiver 38 and would accommodate a low cost, fully integratedamplifier design.

The second calibration mode uses cancellation to determine the relativeamplitude and phase between channels. Referring now to FIG. 5, thesystem represented by 60 sends CW signals 61, 63 from modem 5 throughtwo channels. Using the D/A converter 67, the amplitude and phase of thesignals can be precisely controlled digitally. The combiner's outputsignal level 69 is measured using a directional coupler 71 and adetector 73. When the relative amplitude of the input signals is equaland the phase is 180° apart, the signal amplitude leaving the couplerwill be nearly zero. The energy will be absorbed in the combiner. Usingan iterative search technique, the “cancellation” amplitude and phasevalues can be found very rapidly (10's of trials within milliseconds).

The desired operational amplitude and phase values for normaltransmission have the two signals equal in amplitude and in phase. Thus,the operational amplitude and phase values are the cancellation valueswith an additional 180° phase change.

This processes is repeated with the other channels while keeping onechannel as a reference.

The advantage of this method is that is independent of the receiver 38and would accommodate a low cost, fully integrated amplifier design. Thedisadvantage of this method is that the combiner must be able-todissipate large power levels. However, it is believed that the iterativetechnique will occur so rapidly that little energy will be adsorbed bythe combiner and that that the combiner only need to be able towithstand a high transient voltage (10 W peak power would create about31 volts) and not sustained high power levels.

Referring now to FIG. 6, the third calibration mode is to loop back thesignal 73 from the amplifier 9 to the receiver 38 using the loop backswitch 74 and let the modem measure the amplifier's output signalamplitude and phase relative to the signal input to the amplifier. Thisis illustrated in FIG. 6. The test signal 75 could be a swept CW signal,a special test waveform, or the operational data waveform. Theadditional RF hardware needed for calibration is an RF switch and anattenuator and compared to the previous methods, this approach may havea lower hardware cost. Additionally, the transceiver needs to be able tooperate in a duplex mode (able to simultaneously transmit and receive).However, in many cases the transceiver is not a duplex design and thiscalibration approach would not be applicable.

A variation of the third calibration method is to have pairs of nodes 91calibrate themselves using a communications link loop method as shown inFIG. 7. In this method, the first transceiver 92 transmits a knownwideband test signal and the second transceiver 94 digitizes thewaveform, transforms the signal into sub-bands 95, and determines therelative amplitude and phase numerically. The second transmitter 94 thensends this calibration information 97 to the first transceiver to use toset the amplitude and phase of the sub-bands. A similar technique hasbeen proposed by other researchers to pre-equalize signals to compensatefor multipath effects.

To initiate the process when there is no information on the relativesub-band amplitude and phases, a single sub-band could be used for lowrate communications between the transceivers until the relativeamplitudes and phases are measured.

The communications link loop back approach potentially is the lowestcost calibration method because it requires no additional RF hardwarefor calibration.

An extension of the third calibration technique is to have the modemtransmit calibration signals with different waveforms on the samesub-band channel. FIG. 8B shows each sub-band channel transmitting usingthe same frequency band but using a waveform with different directsequence spreading codes. The receiver 92 or 94 demodulates the fourwaveforms, and determines the relative amplitude and phases between eachchannel. By changing the center frequency of the calibration waveform,the relative amplitude and phase of the channels at any desiredfrequency can be obtained. FIG. 8A shows the waveform in normaloperation with each channel transmitting in different sub-bands.

This extension of the third method calibration of the third method is tointegrate the calibration procedure with the equalizer used by thereceiver 94 to compensate for multipath. Multipath generates similaramplitude and phase variations with frequency as the differences betweenthe RF chains in the sub-band architecture. The receiver 94automatically corrects for the multipath amplitude/phase variations andwould also correct for the RF chain amplitude/phase variations.

To maximize the application of the sub-band transmitter technology to awide range of applications, including defense-related applications, anarchitecture is desired that is independent of the output frequency, thewaveform type, the modem design, and the frequency up conversionprocess. In the embodiment of FIG. 1, the modem 5 splits the signal intosub-bands digitally. This requires that the SBT transmitter beintegrated into the modem or use digital waveform data as an inputsignal. It would be desirable to provide architecture that would beindependent of this requirement. An architecture meeting these goals isshown in FIG. 9 and designated by the reference numeral 100. A universalsub-band transmitter processing module is designated as 101. This moduleenables the sub-band transmitter concept to be generally applied. Theinput signal to the module 101 is an analog IF signal 103 that isdigitized by analog-to-digital converter 105. The digitized signal 107,is split or parsed into sub-bands 109A-109D at 108, calibrated, and isconverted back to analog by converters 111. Four (or more) of theseanalog signals 113A-13D are the universal module output. Each of thesesignals is frequency up converted at 115 and amplified at 117 by RFcomponents that are appropriate for the specific transmitter design.

The universal module 101 uses the cancellation calibration method shownin FIG. 5 and only requires a coupled sample of the output signal. Thisapproach is independent of the output frequency (RF detectors are verybroadband), is independent of the waveform type as long as the A/D andD/A dynamic range are high enough, has no connection to the modem, orthe frequency up conversion. The calibration process could be triggeredby a single control line or could be regularly spaced in time withoutcoordination (the amount of “down time” would be negligible).

The universal module 101 is comprised of an A/D converter 105, FFTfunctions 107, parse and calibration 108, and D/A components 111. Thelargest cost would be for the FPGA or ASIC used for the FFT and otherdigital operations. Depending on the waveform type, the module'shardware cost would be approximately the same as the modem cost becausethey have similar processing loads.

The universal module 101 would be applicable to nearly any widebandwidth radio needing high transmit power (>5 W). There would also bea significant commercial market for cellular base stations and otherhigh power applications.

The inventive sub-band transmitter is novel compared to the technicalapproach used in present high power, low distortion amplifiers.Practically all multi-carrier power amplifier designs are based on thefeed-forward architecture. This architecture, as shown in FIG. 10,consists of two control loops: a signal cancellation loop and an errorcancellation loop. In a feed-forward amplifier, the clean input signal121 is first delayed at 123 and matched in amplitude and phase to theoutput signal 125 of the main power amplifier, and then subtracted fromit at 126. The result is an error signal 127, which contains theinter-modulation products of the main amplifier and only small residuesof the signal itself. This error signal is again amplified at 129 andmatched in delay, amplitude and phase to the output of the mainamplifier 125, and then subtracted from it at 130 to create a cleanoutput signal 131.

Feed-forward is a well-established, solid architecture. It has importantadvantages such as high performance across large bandwidths as well, asunconditional stability. It also has serious drawbacks, which havehistorically limited the economy and performance of the power amplifier.The most serious limitation of the feed-forward architecture is itsrequirement for an extremely accurate RF design. This is needed becauseof the two signal subtractions performed; one in the signal cancellationloop at 126 and one in the error cancellation loop at 130. The accuracyof these subtractions requires that the two signals be matched in delayto fractions of a nanosecond, in amplitude to tenths of a dB, and inphase to tenths of a degree. Typically, this accuracy must be maintainedover 25 MHz of bandwidth, across the entire temperature range of theequipment, for a range of DC voltages, and for a large dynamic range ofthe amplified signal. Providing all this in a very reliable, low cost,mass-produced, high power unit proved to be an extremely difficulttechnological challenge.

The inventive sub-band transmitter concept reduces unintended emissionsby an entirely different method than the feed-forward method. Theinventive sub-band transmitter limits the frequency range of the,distortion signals by limiting the frequency range that each activedevice must amplify.

It should be noted that the both the inventive sub-band transmitter andthe feed-forward techniques can be used together to further improve theamplifier's performance.

The sub-band transmitter of the invention is provides vast improvementsover the prior art by significantly reducing a power amplifier'sundesired out-of-band emissions. A detailed time-domain simulation wasperformed using Matlab that estimated the spurious emission levels ofeach of the three amplifier architectures described in the previoussection. A well-known Taylor series expansion amplifier model was used.This model simulates the non-linear effects of a RF amplifier and themodel was used in the analysis presented in several of the figuresdiscussed below.

The amplifier was modeled using a traditional Taylor Series expansion(see Rohde and Newkirk, Wireless Circuit Design, 91 (2000)). Theexpansion coefficients (k1, k2, and k3) are related to IP₃ and IP₂ asshown below. The IP₂ value was 10 dB larger than the IP₃ value. The IP₃was parametrically varied. The “Taylor Series” amplifier model is

y=k1*x+k2*x.*x+3*x.*x.*x;

where

x = input voltage; y = output voltage; k1 = sqrt (10{circumflex over( )}(gain_dB/10)); linear coefficient k2 = −2*k1{circumflex over( )}2/IP2_V; second order coefficient k3 = −4*k1{circumflex over( )}3/(3*IP₃_V{circumflex over ( )}2); third order coefficient IP3_W =.001*10{circumflex over ( )}(output_IP3_dBm/10); IP3_V = sqrt(2*IP3_W*R); Convert power to voltage IP2_W = .001*10{circumflex over( )}(output_IP2_dBm/10); IP2_V = sqrt (2*IP2_W*R); R = 50 ohms.

Using the model described above, the input signal was a 27 dBm-powerlevel, broadband signal with 1024 equally spaced tones spaced over a 20MHz bandwidth centered at 100 MHz.

FIGS. 11A-C shows the transmitted spectrum with each amplifierarchitectures described previously. FIG. 11A shows the output spectrumusing the system of FIG. 1 wherein each sub-band is amplified. FIG. 11Bshows the output spectrum using the prior art system of FIG. 2 whereinthe entire spectrum is amplified and transmitted (the baseline). FIG.11C shows the system of FIG. 3 wherein four amplifiers operate inparallel on the entire spectrum (baseline with equivalent amplifiers).The spectrum was created using an FFT of the voltage time series with aresolution bandwidth of 61 kHz.

The sub-band transmitter of FIG. 1 produces an output spectrum in FIG.11A with spurious emissions that are greatly reduced (more than 40 dB atmost frequencies) compared to the other architectures as shown in FIGS.11B and 11C. In contrast, there is a large “pedestal” of spuriousemissions surrounding the desired signal in the baseline (FIG. 11B) andalternate four-channel (FIG. 11C) configurations. The spurious signalsare due to cross modulation and typically extend in frequency on bothsides of the desired signal by the signal bandwidth amount. Extensivefiltering would be required to reduce these levels to the required −47.1dBm to −57.1 dBm (in a 61 kHz resolution bandwidth that is currentlymandated as discussed above, see Table 1). By sub-banding, the spectrum“shoulder width” is greatly reduced and no filtering is required (otherthan to remove the harmonic distortion at 200 MHz). It is believed thatby increasing the number of sub-bands from four to eight, that theremaining “shoulder” spurious emissions shown in FIG. 11A would befurther reduced and effectively eliminated.

FIGS. 12A-C show a similar simulation except that a discontinuousspectrum is used. Again, the sub-band transmitter of the inventionprovides very small spurious emissions and even provides a significantreduction in the spectrum level in the spectrum gap (see FIG. 12A). Ifthe emission levels in the gap need further reduction, a tunable notchcould be used. This is much simpler than not employing the inventivetechnology and using a combined tunable bandpass (with variable passband size) and a tunable notch.

As such, an invention has been disclosed in terms of preferredembodiments thereof which fulfills each and every one of the objects ofthe present invention as set forth above and provides new and improvedRF power amplifier architecture.

Of course, various changes, modifications and alterations from theteachings of the present invention may be contemplated by those skilledin the art without departing from the intended spirit and scope thereof.It is intended that the present invention only be limited by the termsof the appended claims.

1. A method of transmitting a signal, comprising: receiving an inputsignal; subdividing the input signal into a plurality of sub-bands, eachsub-band containing a different spectral component of the input signal;up-converting and amplifying each sub-band; and combining the amplifiedsub-bands for transmission.
 2. The method of claim 1, furthercomprising: calibrating the amplified sub-bands so that the combinedsub-bands approximate the signal to be transmitted.
 3. The method ofclaim 2, wherein calibrating the amplified sub-bands further comprises:making an initial relative amplitude and phase measurement for eachsub-band; subsequent to the initial measurement, making amplitude-onlymeasurements for each sub-band; and based on the subsequentmeasurements, determining an amplitude and phase correction for eachsub-band.
 4. The method of claim 3, wherein the amplitude-onlymeasurements are made using a swept CW signal.
 5. The method of claim 2,wherein calibrating the amplified sub-bands further comprises:generating a pair of CW signals; and iteratively varying the relativeamplitudes and phases of the pair of CW signals until a summed output isminimized to determine the relative amplitude and phase values for thesubdividing.
 6. The method of claim 2, wherein calibrating the amplifiedsub-bands further comprises: generating a test signal; amplifying thetest signal; and looping the amplified test signal back to a receiverand modulator for measuring the amplitude and phase of the looped-backsignal relative to the amplitude and phase of a signal corresponding tothe input signal.
 7. The method of claim 6, wherein the test signalcomprises a plurality of test signals, each test signal transmitted overthe same frequency range using different spreading codes.
 8. The methodof claim 2, wherein calibrating the amplified sub-bands furthercomprises: providing a test signal; amplifying the test signal;transmitting the amplified test signal using a first transceiver to asecond transceiver; and determining the relative amplitude and phasevalues of the test signal by digitizing the test signal, transformingthe digitized test signal into sub-bands, and amplifying the sub-bands;wherein the second transceiver transmits the determined relativeamplitude and phase values to the first transceiver to set the amplitudeand phase of the sub-bands for the subdividing step.
 9. The method ofclaim 8, wherein the first receiver includes a loop back switch to allowtransmission of the test signal to the second transceiver and receipt ofthe determined relative amplitude and phase, and the second transceiverincludes a loop back switch to receive the transmitted test signal andto permit transmission of the determined relative amplitude and phase ofthe sub-bands to the first transceiver.
 10. The method of claim 1,wherein the input signal is an analog signal.
 11. A system comprising:an antenna; a processing module to parse a received input signal into aplurality of sub-bands, each sub-band containing a different spectralcomponent of the input signal; an amplifier to amplify at least onesub-band; and a combiner to combine the plurality of sub-bands fortransmission by the antenna.
 12. The system of claim 11, wherein theprocessing module is further configured to calibrate each sub-band. 13.The system of claim 11, further comprising: a plurality of amplifiers,each amplifier configured to amplify one of the plurality of sub-bands.14. The system of claim 11, further comprising a calibration unitconfigured to calibrate amplitude and phase differences for theplurality of sub-bands.
 15. The system of claim 14, wherein thecalibration unit is configured to generate a test signal correspondingto each sub-band and to measure an output signal for each sub-band todetermine amplitude-only corrections.
 16. The system of claim 14,wherein the calibration unit is configured to generate a test signalcorresponding to each sub-band and to measure an output signal for eachsub-band to determine amplitude and phase corrections.
 17. The system ofclaim 14, wherein the calibration unit is configured to generate a pairof test signals and vary the amplitude and phase of the test signals,the system further comprising a detector configured to detect azero-output signal of the combiner to determine the amplitude and phasevalues for the digitized sub-bands.
 18. The system of claim 14,comprising: a loop back switch to receive a test signal from theamplifier; and a first transceiver configured to receive output from theloop back switch and send an output signal to the processing module,wherein the processing module is configured to measure the amplitude andphase of the output signal received from the transceiver relative to thetest signal to determine amplitude and phase corrections.
 19. The systemof claim 18, further comprising a second transceiver, the secondtransceiver configured to: receive the test signal from the firsttransceiver; transform the test signal into sub-bands; determine therelative amplitude and phases of the sub-bands; and transmit thedetermined amplitude and phases to the first transmitter system.